1. Field of the Invention
The present invention relates to a flip-flop circuit (called “FF” below) using a semiconductor device that achieves an increase in operating speed and a reduction in power consumption and to a frequency divider using the flip-flop circuit.
2. Description of the Related Art
Conventionally, as technology relating to a frequency divider that achieves an increase in operating speed and a reduction in power consumption, there has been the frequency divider described in the following document, for example.
N. Krishnapura and Peter R. Kingget, “A 5.3 GHz Programmable Divider for HiPerLAN in 0.25 μm CMOS,” IEEE Journal of Solid-State Circuits, Vol. 35, No. 7, July 2000, pp. 1019-1024.
In recent years, LSI have become increasingly integrated and sophisticated, and the applied fields thereof have expanded in a wide range. Particularly in accompaniment with the development of wireless communication systems, technological demands to reduce the power consumption of LSI and for high-speed operation of a GHz order have become stronger over the years. Among those, reducing the power consumption of PLL (Phased Locked Loop), which is an important configural circuit in wireless communication systems, has become the most critical issue. And in particular, reducing the power consumption of frequency dividers is extremely effective with respect to reducing the power consumption of PLL. The reason is because a frequency divider used in PLL requires operation of a GHz order depending on the specifications. A frequency divider that operates on a GHz order is an elemental circuit that consumes the most power, and reducing the consumption of this circuit is linked to reducing the power consumption of PLL. A configuration that uses MOS Current Mode Logic (called “MCML” below) has been proposed as a frequency divider that enables a reduction in power consumption while satisfying high-speed operation of a GHz order of a frequency divider. A paper on an applied example of a PLL-use frequency divider using MCML is described in the aforementioned document.
FIG. 7 is a circuit diagram showing a toggle FF (called “TFF” below) 10 using the conventional MCML technology described in the aforementioned document. FIG. 8 is a configural diagram showing a conventional ½ N frequency divider configured as a result of N stages of the TFF 10 of FIG. 7 being cascade-connected.
The MCML TFF 10 of FIG. 7, which is the basic configural circuit of the ½ N frequency divider shown in FIG. 8, is configured by: P-channel MOS transistors (called “PMOS” below; the ON resistance is R and the ON current is I) 11-1 to 11-4 whose input/output signals (e.g., a clock ck that is a first input pulse, an inversion clock ckb that is a second input pulse, a signal of a first output terminal out, and a signal of an inversion output terminal outb that is a second output terminal) all comprise differential complementary signals and which are first, second, third and fourth load elements for obtaining signal amplitude; and first and second latch portions 12A and 12B that comprise N-channel MOS transistors (called “NMOS” below) configuring logic.
The first and second latch portions 12A and 12B are connected to the output terminal out and the inversion output terminal outb. The output terminal out is connected to a first power supply potential VDD via a drain electrode/source electrode of the PMOS 11-1 of the first latch portion 12A, and a gate electrode of the PMOS 11-1 is connected to a ground potential VSS that is a second power supply potential. The inversion output terminal outb is connected to the power supply potential VDD via a drain electrode/source electrode of the PMOS 11-2, and a gate electrode of the PMOS 11-2 is connected to the ground potential VSS. A first output node m1 of complementary first and second output nodes m1 and m2 of the second latch portion 12B is connected to the power supply potential VDD via a drain electrode/source electrode of the PMOS 11-3, and a gate electrode of the PMOS 11-3 is connected to the ground potential VSS. The second output node m2 is connected to the power supply potential VDD via a drain electrode/source electrode of the PMOS 11-4, and a gate electrode of the PMOS 11-4 is connected to the ground potential VSS.
The first latch portion 12A is configured by first to eighth NMOS 12-1 to 12-8. The output terminal out is connected to the ground potential VSS via a drain electrode/source electrode of the first NMOS 12-1, a first node n1, and a drain electrode/source electrode of the second NMOS 12-2, and is connected to the ground potential VSS via a drain electrode/source electrode of the third NMOS 12-3, a second node n2, and a drain electrode/source electrode of the fourth NMOS 12-4. A gate electrode of the NMOS 12-1 is connected to the inversion output terminal outb, the inversion clock ckb is applied to a gate electrode of the NMOS 12-2, a gate electrode of the NMOS 12-3 is connected to the second output node m2, and the clock ck is applied to a gate electrode of the NMOS 12-4.
The inversion output terminal outb is connected to the ground potential VSS via a drain electrode/source electrode of the fifth NMOS 12-5, a third node n3, and a drain electrode/source electrode of the sixth NMOS 12-6, and is connected to the ground potential VSS via a drain electrode/source electrode of the seventh NMOS 12-7, a fourth node n4, and a drain electrode/source electrode of the eighth NMOS 12-8. A gate electrode of the NMOS 12-5 is connected to the output terminal out, the inversion clock ckb is applied to a gate electrode of the NMOS 12-6, a gate electrode of the NMOS 12-7 is connected to the first output node m1, and the clock ck is applied to a gate electrode of the NMOS 12-8.
The second latch portion 12B is configured by ninth to sixteenth NMOS 12-9 to 12-16. The first output node m1 is connected to the ground potential VSS via a drain electrode/source electrode of the ninth NMOS 12-9, a fifth node n5, and a drain electrode/source electrode of the tenth NMOS 12-10, and is connected to the ground potential VSS via a drain electrode/source electrode of the eleventh NMOS 12-11, a sixth node n6, and a drain electrode/source electrode of the twelfth NMOS 12-12. A gate electrode of the NMOS 12-9 is connected to the second output node m2, the clock ck is applied to a gate electrode of the NMOS 12-10, a gate electrode of the NMOS 12-11 is connected to the output terminal out, and the inversion clock signal ckb is applied to a gate electrode of the NMOS 12-12.
The second output node m2 is connected to the ground potential VSS via a drain electrode/source electrode of the thirteenth NMOS 12-13, a seventh node n7, and a drain electrode/source electrode of the fourteenth NMOS 12-14, and is connected to the ground potential VSS via a drain electrode/source electrode of the fifteenth NMOS 12-15, an eighth node n8, and a drain electrode/source electrode of the sixteenth NMOS 12-16. A gate electrode of the NMOS 12-13 is connected to the output node m1, the clock ck is applied to a gate electrode of the NMOS 12-14, a gate electrode of the NMOS 12-15 is connected to the inversion output terminal outb, and the inversion clock ckb is applied to a gate electrode of the NMOS 12-16.
As for the operation of the MCML TFF 10 of FIG. 7, the PMOS 11-1 to 11-4 are always in an ON state, and in an initial state, when the output terminal out is “0”, the inversion output terminal outb is “1”, the output node m1 is “1” and the output node m2 is “0”, for example, then the NMOS 12-1, 12-7, 12-13 and 12-15 are ON and the NMOS 12-3, 12-5, 12-11 and 12-9 are OFF.
When the clock ck becomes “1” and the inversion clock ckb becomes “0”, then the NMOS 12-4, 12-8, 12-10 and 12-14 are switched to an ON state and the NMOS 12-2, 12-6, 12-12 and 12-16 are switched to an OFF state. Then, the inversion output terminal outb is lowered toward the ground potential VSS by the NMOS 12-7 and 12-8 in the ON state and becomes “0”. At this time, the output node m2 is maintained at “0” and the output node m1 is maintained at “1” by the NMOS 12-13 and 12-14 in the ON state. When the inversion output terminal outb becomes “0”, then the NMOS 12-1 is switched to an OFF state, and the output terminal out is raised in the power supply potential VDD direction via the PMOS 11-1 and becomes “1”. Thus, the output terminal out and the inversion output terminal outb are inverted from “0” and “1” to “1” and “0”.
When the clock ck becomes “0” and the inversion clock ckb becomes “1”, then the NMOS 12-4, 12-8, 12-10 and 12-14 are switched to an OFF state and the NMOS 12-2, 12-6, 12-12 and 12-16 are switched to an ON state. Then, the inversion output terminal outb is maintained at “0” and the output terminal out is maintained at “1” by the NMOS 12-5 and 12-6 in the ON state. At this time, the output node m1 is lowered toward the ground potential VSS by the NMOS 12-11 and 12-12 in the ON state and becomes “0”, and the output node m2 is raised toward the power supply potential VDD via the PMOS 11-4 by the NMOS 12-13 and 12-14 in the OFF state and becomes “1”.
Next, when the clock ck becomes “1” and the inversion clock ckb becomes “0”, the output terminal out is lowered toward the ground potential VSS by the NMOS 12-3 and 12-4 in the ON state and becomes “0”, and the inversion output terminal outb is raised toward the power supply potential VDD by the PMOS 11-2 and becomes “1”. Thus, the NMOS 12-11 is switched to an OFF state, the NMOS 12-15 is switched to an ON state, the output node m1 is raised toward the power supply potential VDD via the PMOS 11-3 and becomes “1”, and the output node m2 is lowered toward the ground potential VSS by the NMOS 12-13 and 12-14 in the ON state and becomes “0”. Thus, the output terminal out and the inversion output terminal outb are inverted from “1” and “0” to “0” and “1”.
In this manner, in the MCML TFF 10 of FIG. 7, each time the clock ck becomes “1” and the inversion clock ckb becomes “0”, the logic levels of the output terminal out and the inversion output terminal outb are inverted, and counting is performed to halve the number of inputted clocks ck (or inversion clocks ckb). For this reason, when N stages of the TFF 10 are cascade-connected as in the frequency divider of FIG. 8, the frequency of the inputted clock ck (or inversion clock ckb) becomes divided into ½ N.
Here, in the MCML TFF 10 of FIG. 7, a high level (called “H level” below) of signal is defined as the power supply potential VDD level and a low level (called “L level” below) is defined as a level that has fallen IR from the power supply potential VDD level, so the signal amplitude becomes IR.
Whereas the signal amplitude of a commonly used complementary MOS (called “CMOS” below) logic circuit is the power supply potential VDD, the signal amplitude of an MCML TFF 10 becomes IR, and this is linked to a reduction in charge and discharge time and means an improvement in operating speed. Moreover, in contrast to a CMOS logic circuit, there is no logic threshold voltage with respect to changes in the input signal, and the changing of the output signal by the input signal amplitude of about the threshold voltage of NMOS also contributes to increasing speed. Additionally, because the MCML TFF 10 operates by differential complementary signals, being resistant to common mode noise also enables high-speed operation.
From the standpoint of power consumption, in the MCML TFF 10, in contrast to a CMOS logic circuit, current always flows from the power supply potential VDD because the PMOS 11-1 to 11-4 that are loads are ON. Because the MCML TFF 10 operates at a higher speed than a CMOS logic circuit if its operating frequency is in a high range, the power supply potential VDD can be reduced, and a reduction in power consumption is possible in comparison to a CMOS circuit. In contrast, a CMOS logic circuit does not in principle consume power when the circuit is not operating because either the PMOS or the NMOS are always OFF. However, in the MCML TFF 10, current always flows from the power supply potential VDD, so that when the circuit is not operating or when the input signal frequency of the TFF 10 is slow and the operating rate of the gate is extremely low, less power is consumed when a CMOS logic circuit is used. For this reason, the MCML TFF 10 and a frequency divider using the MCML TFF 10 can be said to be circuits directed toward a GHz order.
Because the MCML TFF 10 and a frequency divider using the MCML TFF 10 operate at a high speed in comparison to when a CMOS logic circuit is used, it becomes possible to reduce the power supply potential VDD and they are suited to reducing power consumption at the time of high-speed operation at a GHz order. However, in a range where the operating frequency is low, power consumption becomes greater in comparison to a CMOS logic circuit. Because the frequency of the signal falls toward the TFF of the later stages, a method can also be considered where the MCML TFF 10 is used when it is necessary to cause the frequency divider to operate at a high speed and where a CMOS logic circuit is used when the frequency has fallen a certain extent. However, the great advantage of the MCML TFF 10 that a CMOS logic circuit does not have is that it is resistant to common mode noise because it operates by differential complementary signals. This characteristic becomes an extremely important advantage when stable operation of the frequency divider with respect to reducing the voltage of the power supply potential VDD is considered. For that reason, the MCML TFF 10 and a frequency divider using the MCML TFF 10 become extremely important circuits with respect to reducing the voltage of the power supply potential VDD in the future. Thus, reducing power consumption while preserving the differential circuit format having the characteristic that it is resistant to common mode noise becomes an issue of the MCML TFF 10 and a frequency divider using the MCML TFF 10.
As a solution therefor, when, for example, a high-frequency circuit block and a low-frequency circuit block in a frequency divider are configured using the conventional MCML TFF 10, a reduction in power consumption can be achieved when the dimensions of the PMOS 11-1 to 11-4 that are load transistors in the low-frequency circuit block are made much smaller than the dimensions of the PMOS 11-1 to 11-4 that are load transistors in the high-frequency circuit block, but on the other hand, the potential arises for the rise speed of the H level signal from the TFF 10 of the low-frequency circuit block to fall so that, as a result, the desired circuit operation can no longer be achieved. Conversely, when the dimensions of the PMOS 11-1 to 11-4 in the low-frequency circuit block are made much smaller than the dimensions of the PMOS 11-1 to 11-4 in the high-frequency circuit block, the rise speed of the H level signal from the TFF 10 of the low-frequency circuit block can be maintained at a desired level, but on the other hand, it becomes difficult to achieve a reduction in power consumption.
Consequently, there has been a strong desire to achieve an FF and a frequency divider using the FF that can reduce power consumption while maintaining the rise speed of the H level signal (or the fall speed of the L level signal) from the TFF.